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A New Compact Microstrip-Fed Dual-Band Coplanar Antenna for WLAN Applications
Rohith K. Raj, Manoj Joseph, C. K. Aanandan, K. Vasudevan, Senior Member, IEEE, and P. Mohanan, Senior Member, IEEE

Abstract—A novel compact microstrip fed dual-band coplanar antenna for wireless local area network is presented. The antenna comprises of a rectangular center strip and two lateral strips miprinted on a dielectric substrate and excited using a 50 crostrip transmission line. The antenna generates two separate resonant modes to cover 2.4/5.2/5.8 GHz WLAN bands. Lower resonant mode of the antenna has an impedance bandwidth (2:1 VSWR) of 330 MHz (2190–2520 MHz), which easily covers the required bandwidth of the 2.4 GHz WLAN, and the upper resonant mode has a bandwidth of 1.23 GHz (4849–6070 MHz), covering 5.2/5.8 GHz WLAN bands. The proposed antenna occupy an area of 217 mm2 when printed on FR4 substrate . A rigorous experimental study has been conducted to confirm the characteristics of the antenna. Design equations for the proposed antenna are also developed.

( = 4 7)

Index Terms—Coplanar waveguide, dual-band antennas, printed antennas, wireless local area networks (WLANs).

I. INTRODUCTION IRELESS LOCAL area networks (WLAN) are being widely recognized as a viable, cost effective and high speed data connectivity solution, enabling user mobility. The rapid developments in WLAN technologies demand the integration of IEEE 802.11 WLAN standards of the 2.4 GHz (2400–2484 MHz), 5.2 GHz (5150–5350 MHz) and 5.8 GHz (5725–5825 MHz) bands into a single unit. To comply with the above requirements, compact high performance multiband antennas with excellent radiation characteristics are required. Various kinds of antennas, such as a multiband inverted-L monopole with meandered wire and a conducting triangular section for dual band operation [1] and a microstrip fed printed double-T monopole antenna [2] were reported to provide dual band characteristics to cover the 2.4/5.2 and 5.8 GHz WLAN bands. Wu et al. [3] reported a dual broadband slot antenna, in which the two wide resonances were obtained by using a U-shaped strip inset at the center of the slot antenna of dimensions 75 mm 75 mm, on a substrate of relative permittivity 4.7 and thickness 0.8 mm. The dual-band WLAN dipole antenna proposed by Zhijun Zhang et al. [4], fabricated on FR4 substrate with dimensions 12 mm 45 mm uses an internal matching circuit to completely cover the three WLAN bands.
Manuscript received October 3, 2005; revised July 20, 2006. The authors are with the Centre for Research in Electromagnetics and Antennas (CREMA), Department of Electronics, Cochin University of Science and Technology, Cochin 682-022, Kerala, India (e-mail: drmohan@cusat.ac.in). Color versions of Figs. 2, 3 and 11 are available online at http://ieeexplore. ieee.org. Digital Object Identifier 10.1109/TAP.2006.886505

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There are several papers on dual band monopole antennas with parasitic strips. Huang et al. [5] proposed a microstrip fed dual-band monopole antenna comprising of a C-shaped radiating element and a parasitic radiating strip directly shorted to the ground plane. The monopole with shorted parasitic inverted-L [6] and dual-band flat plate antenna with a shorted parasitic element for laptop applications [7] were also proposed to operate in 2.4 and 5 GHz WLAN bands. In these designs the longer strip controls the lower band of the antenna, while the shorter strip and the parasitic strip together generate the wide operating band for the upper resonance. In all the above designs the wide bandwidth is achieved by shorting the monopole or the parasitic patches. However the shorting will reduce the overall gain of the antenna. The wideband coplanar waveguide fed monopole antenna with two parasitic elements [8] utilizes the electromagnetic coupling effect for achieving 3.1–11 GHz operation. But the overall size of the antenna is large. Moreover, the antenna resonant frequencies depend on more parameters like slot width, length etc. This may lead to design complexities. In this paper we are presenting a compact dual-band planar antenna without any parasitic elements. The design is akin to a coplanar transmission line excited asymmetrically. In conventional coplanar waveguide designs width of the center strip is always kept small and thus surface current orthogonal to the direction of power flow is negligible. Shigesawa et al. have shown [9], [10] that the dominant mode on conventional coplanar waveguide becomes leaky instead of purely bound only when the frequency of operation is sufficiently high. This power leakage occurs only in the form of surface waves. It is apparent that this behavior cannot be used to design compact efficient radiators, because the power leakage from the coplanar waveguide structure starts only at higher frequency bands. Tsuji et al. [11] presented a detailed study on the power leakage from a coplanar wave guide. They concluded that width of the lateral strips (ground strips) plays a strong role in the leakage properties, whereas width of the center strip and gaps play only a secondary role. In this work a compact coplanar waveguide structure is demonstrated as an efficient radiator at lower microwave frequency band by optimizing various dimensions of the waveguide and feed point. The new antenna design approach does not utilize the surface wave modes excited on the coplanar wave guide structure, but uses the radiation from the two gaps of the coplanar structure. The optimum design provides good impedance bandwidth, gain and nearly omni directional radiation pattern. The proposed antenna is termed as coplanar antenna due to the similarity of the radiating structure to a conventional coplanar waveguide. The antenna consists of three rectangular strips printed on a dielectric substrate. A 50

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TABLE I IE3D SIMULATION RESULTS SHOWING THE RESONANCE BEHAVIOR OF THE COPLANAR WAVE GUIDE l = 15mm, W = 15mm, g = 0:3mm, WHEN PRINTED ON FR4 SUBSTRATE HAVING " = 4:7 AND THICKNESS h = 1:6mm

Fig. 1. Coplanar wave guide with finite lateral strips and the feed point P =P when printed on a substrate of thickness h and relative permittivity " . (a) Top view. (b) Cross-section view.

microstrip feed line is employed to excite the antenna with two wide resonant modes. The two lateral patches are shorted to the ground plane of the microstrip line using vias. A prototype of the antenna was designed, fabricated and characteristics were experimentally verified. The antenna has a dimension of 24.1 and mm 9 mm when printed on FR4 substrate with thickness 1.6 mm. The radiation mechanism, design procedure and experimental results of the antenna are presented in the following sections. II. RESONANCE AND RADIATION PHENOMENA A detailed simulation has been carried out using method of moment based Zeland IE3D electromagnetic solver to bring out the resonant conditions of the coplanar waveguide structure. In order to study the phenomenon, an open circuited section of finite coplanar waveguide as shown in Fig. 1 with characteristic mm, impedance 50 (in this case, center strip width mm, mm, length lateral strip width mm, on FR4 substrate with , thickness mm) is selected and its frequency versus return loss characteristics over a wide frequency range (1–10 GHz) is observed. values, at difThe process has been repeated for different ferent feed locations along the width of center strip. The characteristics of the structure when the feed point is at center and when the feed point is shifted to the corner , as shown in Fig. 1, have been illustrated in Table I. When feed point is at the center, the structure behaves as an open circuited coplanar wave guide transmission line. In this case the return loss is 0dB, indicating high reflection from the input port due to the open circuit at the other end. But when the feed location is at the corner, good input matching is observed at a particular frequency (10 % centered at the frequency as indicated dB band width is in Table I), indicating a resonance on the coplanar waveguide structure. The return loss value in this case only indicates the fact that the coplanar waveguide structure is matched even when the other end is open circuited. But it doesn’t convey any hint about the power leakage from the structure in the form of radiation. The radiation from the structure for this mode is confirmed by calculating the gain of the structure. The gain of this structure obtained from IE3D simulation for mm is 1.2 dBi at 4.825 GHz, indicates the possibility for obtaining good

Fig. 2. Simulated current distribution of (a) conventional coplanar wave guide and (b) coplanar waveguide with feed point at the corner. w = 3 mm, W = 15 mm, l = 15 mm, g = 0:3; " = 4:7, and thickness h = 1:6 mm.

radiation from the structure, when the other structural parameters are properly optimized. It is also noted that the resonant increases. Thus Table I obfrequency decreases as the width viously shows that the structure excites a resonant mode with good input matching only when the feed point is away from the center of the signal strip, along . Fig. 2 shows the simulated current distribution using IE3D for both cases. It is obvious from Fig. 2(a) that the transverse variation of surface current is negligible along the width for a conventional coplanar wave guide. But when feed point is moved to the corner, a transverse surface current appears along the width of the signal strip (at the feeding edge) in addition to its longitudinal variations (along the length) on the wave guide, as shown in Fig. 2(b). The “U” shaped resonant path thus formed on the center strip is half wave in length, excites resonance on the coplanar wave guide structure. The important conditions brought out from the study to make the coplanar wave guide structure as an efficient compact radiator are as follows. 1. A coplanar waveguide is able to excite a resonant mode at a lower frequency band on the structure when the feed point

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Fig. 3. Simulated current distribution of a coplanar wave guide with wide center strip and feed points at the extreme corners of the strips. w = 9 mm, W = 5 mm, l = 9 mm, g = 0:8; " = 4:7, and thickness h = 1:6 mm.

Fig. 4. Geometry of the proposed dual-band microstrip line fed compact coplanar antenna printed on a dielectric substrate. (a) Cross sectional view. (b) Top view.

is away from the center of the signal strip, along the width (usually at the corner). 2. Length “ ” of the waveguide should be less than , to is the wave obtain a dipole like radiation pattern, where length in the dielectric corresponding to the desired frequency of operation. 3. The ratio between width of the lateral strip and the signal is not very critical, but the ratio 1:2.5 is suitstrip able for obtaining good 2:1 VSWR band width and broad is large a second radiation coverage. Moreover, when resonant mode will be excited on the structure. 4. The ground connection to the lateral strips can be chosen to be at the extreme corners of the two strips to excite the lowest possible mode on the structure. In this case the current variation on the lateral strips should also be taken into account to calculate the resonant frequency. The electromagnetic coupling of center strip to the lateral strips through the gaps excites a full cycle variation for this mode on the structure. Fig. 3 shows the simulated current distribution on a coplanar wave guide structure having wide center strip width with feed points at the extreme corners. The “U” shaped current path on the center strip and the “L” and “reflected L” shaped current path on the lateral strips due to the electromagnetic coupling are clear from the figure.

III. COPLANAR ANTENNA DESIGN Fig. 4 shows the geometry of the proposed dual-band coplanar antenna. The antenna comprises of three rectangular conducting strips; a center strip and two similar lateral strips, printed on one side of a dielectric substrate. The three rectangular strips are of equal and for the center strip and lateral lengths and widths strips respectively. The gap g separates the center strip and the two lateral conductors. A 50 microstrip feed line of width with ground plane dimensions of length and width is connected to the side EF of the center strip as shown in Fig. 4.

Position of the feed line is fixed at the end of side EF to excite the two resonant modes of the proposed antenna with good impedance matching. In order to attain maximum compactness, the extreme points A and J of the lateral strips are connected to the ground plane of the microstripline using vias via and via respectively. Position of vias on the two lateral strips has strong influence on the resonant frequency. The ground plane is truncated underneath the three rectangles as shown in Fig. 4. When antenna is energized, two distinct resonant modes are excited, which generate two wide bands with orthogonal polarof the ization characteristics. The lower resonant frequency antenna is due to a “U” shaped resonant path on the center strip, utilizing the sides HE, EF, and FG, which is equivalent to a half wavelength in substrate and the “L” and “reflected L” shaped resonant path on the two lateral strips utilizing the sides LI, IJ and CB, BA for the right and left lateral strips in Fig. 4, respectively. The current variation on each lateral strip has quarter wave in length. Thus the surface current on the structure at the lower resonant frequency need a full cycle variation. That is ; where is the wavelength corresponding to lower resonant frequency in the substrate. The in the expression is due to the length of the two vias term equal to the substrate height. The gap should be small compared to the wavelength corresponding to the lower resonance to obtain good electromagnetic coupling between the center conis ductor and lateral strips. The upper resonant frequency of the center strip, corresponding obtained due to the width to a half wave length variation in substrate. The radiation of the antenna at the lower resonant band is due to the fields along the length at the two gaps separating center strips from the two lateral strips. This field is due to the electromagnetic coupling between the center strip and the two lateral strips. In other words, the antenna at the lower resonant frequencies can be described as an array of two identical sources separated by a distance , excited in phase. This will produce broad side radiation. In order to maintain a nearly constant phase of the electric field at the two gaps, the dimension should be . The simulation result shows that more fields less than exist between the side EF of the center strip and the ground plane of the microstrip line at the upper resonant mode of the

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proposed antenna. This fringing field contributes radiation at the upper resonant frequency. Thus the radiation mechanism can be approximated to the radiation from a slot lying on the plane of the antenna in the coordinate system indicated in Fig. 4. The microstrip transmission line is the best suited feed for this antenna configuration due to its bottom ground layer, which is required to excite the second resonant mode of the antenna. The following design procedure can be used to design this antenna with good radiation characteristics. The design procedure is as follows. 1. Select any substrate with relative dielectric constant and thickness , and calculate the width of the microstrip transmission line for 50 characteristic impedance. using the fol2. Calculate width of the center strip lowing:
Fig. 5. Measured and simulated return loss for the proposed compact microstrip fed dual band coplanar antenna for 2.4/5.2/5.8 GHz WLAN application.

(1) where is the velocity of light. Since the field components are not confined to the substrate has to be used in alone the effective dielectric constant calculations instead of relative permittivity of the substrate The coefficients 0.12 and 0.98 were derived empirically after studying the effect of ground plane on the two resonant frequencies. 7. The two extreme corners of the lateral conductors are connected to ground plane of the microstrip line using vias or conducting pins. IV. RESULTS AND DISCUSSION 3. The length of the three rectangular strips is then calculated as The dimensions of the proposed antenna intended to operate in 2.4/5.2/5.8 GHz WLAN bands, calculated using the (1)–(7) , loss tangent 0.02 and thickness for FR4 substrate with mm are mm, mm, mm, mm, ground plane dimensions mm and mm. The lower band covers 2.4 GHz WLAN, whereas the upper band covers 5.2/5.8 GHz WLAN bands. A prototype of the antenna is constructed and tested. Measurements were carried out using HP8510C vector network analyzer. Fig. 5 shows the measured and simulated return loss characteristics of the antenna. The transmission line matrix (TLM) based microstripes simulation package was employed for the simulation. Good agreement between measured values and simulation is obtained. From the dB measured results, the lower bandwidth determined by return loss reaches 330 MHz (2190–2520 MHz, about 14% centered at 2.34 GHz) and covers the 2.4 GHz WLAN band, which is greater than the required bandwidth (84 MHz) for the 2.4 GHz WLAN band. The measured bandwidth for the upper band is 1.23 GHz (4849–6070 MHz, about 22% centered at 5.26 GHz) and covers the two WLAN bands (5150–5825 MHz) easily. The wide impedance bandwidths obtained for the two bands allows manufacturing tolerance during fabrication of the antenna. The large impedance bandwidth in the two bands is due to the increased surface area of the strips. The antenna on RT Duroid and mm) offers 11% bandwidth in substrate ( the lower band and 21% in the higher band, whereas on Alumina and mm) the antenna offers 10% bandwidth in the lower band and 20% in the higher band. Effects of var, and of the proposed antenna ious dimensions such as on FR4 substrate are also studied to confirm the two modes of the antenna.

(2)

(3) 4. Width of the lateral conductors following: is calculated using the

(4) where is the thickness of the dielectric substrate. 5. Gap separating center strip from the lateral strips is then calculated

(5) The coefficients 0.15 and 0.014 in (3) and (5), respectively, were obtained after exhaustive experimental and simulation studies. These values were arrived at from curve fitting technique. 6. The ground plane dimensions are calculated using the following:

(6) (7)

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TABLE II CHARACTERISTICS OF PROPOSED DUAL BAND COPLANAR ANTENNA WITH VARIOUS LENGTHS (l) AND A FIXED W = 12:5mm, W = 5mm, g = 0:8mm WITH GROUND PLANE DIMENSION L = 10mm AND W = 50mm WHEN PRINTED ON FR4 SUBSTRATE HAVING " = 4:7; Loss Tangent = 0:02 AND THICKNESS h = 1:6mm

TABLE III CHARACTERISTICS OF PROPOSED DUAL BAND COPLANAR ANTENNA WITH VARIOUS WIDTHS (W ) AND A FIXED l = 9mm W = 5mm, g = 0:8mm WITH GROUND PLANE DIMENSION L = 10mm AND W = 50mm WHEN PRINTED ON FR4 SUBSTRATE HAVING " = 4:7; Loss Tangent = 0:02 AND THICKNESS h = 1:6mm

TABLE IV CHARACTERISTICS OF PROPOSED DUAL BAND COPLANAR ANTENNA WITH VARIOUS WIDTHS (W ) AND A FIXED l = 9mm W = 12:5mm, g = 0:8mm WITH GROUND PLANE DIMENSION L = 10mm AND W = 50mm WHEN PRINTED ON FR4 SUBSTRATE HAVING " = 4:7; Loss Tangent = 0:02 AND THICKNESS h = 1:6mm

A. Effects of Antenna Length l Table II illustrate the variation in resonant frequency, return loss and percent bandwidth in the two bands as changes. 10, 11, and 13 mm. Three cases have been studied, i.e., Here, all other parameters of the antenna are kept same as that in the previous design. It can be seen that the lower resonance decreases as increases and the upper resonance remains unchanged. Because as increases symmetrically for the three strips, the resonant length for the first band increases four times and thus the resonant frequency in the first band decreases more rapidly. It is also noted that the bandwidth of the antenna’s second resonant band increases slightly as increases. This is due to the increase in surface area of the center strip when increases. B. Effects of Center Strip Width As explained in the previous section, the upper resonance , corresponding to of the antenna is due to the dimension
Fig. 6. Measured and simulated E and H plane radiation patterns of the proposed compact microstrip-fed dual-band coplanar antenna on FR4 substrate at 2.4 GHz.

a half wavelength variation of the surface current. This can be confirmed by studying the variation in resonant frequency when changes. Table III shows the variation of antenna characterinistics with different values of . The table shows that as creases, the upper resonance decreases. It affects the lower band increases, as explained, the resonant path also. Because as for lower resonance increases and thus decreases. It can be seen that the bandwidth in the first band increases slightly as increases. Because the increase in increases the surface area for the surface current on the center strip at the lower resonant frequency, where as it doesn’t increase the surface area for the current on the strip corresponding to upper resonance, and thus the bandwidth remains unchanged in the second band.

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Fig. 7. Measured and simulated E and H plane radiation patterns of the proposed compact microstrip-fed dual-band coplanar antenna on FR4 substrate at 5.2 GHz.

Fig. 8. Measured and simulated E and H plane radiation patterns for the proposed compact microstrip-fed dual-band coplanar antenna on FR4 substrate at 5.8 GHz.

C. Effects of Lateral Strip Width Table IV depicts the effect of on the resonant frequencies of the proposed antenna and its characteristics. increases symmetrically for the two latIt shows that as eral strips the resonant frequency for the first band decreases, inwhere as the second band remains unchanged. When creases symmetrically for the two strips, the resonant length for the first band increases two times, and thus the resonant frequency for the first band decreases. The experimental observations conclude that the lower resonance is obtained by a full wavelength variation of surface current on the antenna structure

and upper resonance is due to the half wavelength variation of the surface current along the width of the center strip. The measured and simulated principal plane radiation patterns of the proposed WLAN antenna at 2.4, 5.2, and 5.8 GHz are shown in Figs. 6–8, respectively. As expected, the radiation pattern at 2.4 GHz has figure of eight shape in the E plane and is found to be nearly non directive in the H plane. The antenna has the same radiation pattern throughout the 2:1 VSWR bandwidth in the first band. Radiation patterns at 5.2 and 5.8 GHz are found to be slightly distorted. This is due to the nonuniform distribution of the field in the slot approximating the fringing field from the side EF to the ground plane. Large cross polarization levels are observed in the radiation patterns. It is found that

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Fig. 11. Photograph of the proposed compact microstrip-fed dual-band coplanar antenna designed for WLAN applications.

Fig. 9. Measured and simulated gain of the proposed dual-band coplanar antenna on FR4 substrate in 5.2 GHz WLAN band.

structure, obtained by suitably selecting the feed point on a wide signal strip. The generalized design procedure for the antenna for any two bands of operation has also been developed. The compactness of the antenna together with the excellent radiation characteristics suggests its direct use in PCMCIA WLAN cards, Bluetooth enabled gadgets, etc. ACKNOWLEDGMENT The authors thankfully acknowledge Prof. K. G. Nair, Cochin University of Science and Technology, for his constant encouragement, suggestions and advice. The authors are also grateful to UGC, Government of India, Kerala State Council for Science Technology and Environment (KSCSTE) and the Defence Research Development Organization (DRDO), Government of India for providing financial assistance. REFERENCES

Fig. 10. Measured and simulated gain of the proposed dual-band coplanar antenna on FR4 substrate in 5.8 GHz WLAN band.

the fringing field from EF to ground plane is responsible for high cross polarization along -direction in the lower band. Similarly the fringing field between the lateral strips and the center strip is responsible for high cross polarization component along -direction in the higher band. The simulated and measured gains dBi and across the 2.4 GHz WLAN band are dBi, respectively. Figs. 9 and 10 shows the measured and simulated gains against frequency of the antenna in the 5.2 and 5.8 GHz WLAN bands. The maximum gains obtained in 5.2 and 5.8 GHz WLAN bands are 3 and 4.2 dBi, respectively. Efficiency of the antenna designed on FR4 substrate is estimated to be 73% at 5.2 GHz, whereas the estimated efficiency of the similar antenna on RT Duroid substrate is 92% at 5.2 GHz. A photograph of the proposed antenna when printed on FR4 substrate is shown in Fig. 11. V. CONCLUSION A novel compact dual band coplanar antenna suitable to completely cover the IEEE 802.11 WLAN standards of 2.4, 5.2 and 5.8 GHz WLAN bands has been demonstrated. The antenna uses two resonant modes excited on a coplanar wave guide

[1] H.-D. Chen, J.-S. Chen, and Y.-T. Cheng, “Modified inverted-L monopole antenna for 2.4/5 GHz dual-band operations,” IEE Electron. Lett., vol. 39, no. 22, Oct. 2003. [2] Y.-L. Kuo and K.-L. Wong, “Printed double-T monopole antenna for 2.4/5.2 GHz dual band WLAN operations,” IEEE Trans. Antennas Propag., vol. 51, no. 9, pp. 2187–2192, Sep. 2003. [3] J. W. Wu, H. M. Hsiao, J. H. Lu, and S. H. Chang, “Dual broadband design of rectangular slot antenna for 2.4 and 5 GHz wireless,” IEE Electron. Lett., vol. 40, no. 23, Nov. 2004. [4] Z. Zhang, M. F. Iskander, J.-C. Langer, and J. Mathews, “Dual-band WLAN dipole antenna using an internal matching circuit,” IEEE Trans. Antennas Propag., vol. 53, no. 5, May 2005. [5] C. Y. Huang and P. Y. Chiu, “Dual-band monopole antenna with shorted parasitic element,” IEE Electron Lett., vol. 41, no. 21, pp. 1154–1155, Oct. 2005. [6] J.-Y. Jan and L.-C. Tseng, “Small planar monopole antenna with a shorted parasitic inverted-L wire for wireless communications in the 2.4, 5.2 and 5.8 GHz bands,” IEEE Trans. Antennas Propag., vol. 52, no. 7, pp. 1903–1905, Jul. 2004. [7] K.-L. Wong, L.-C. Chou, and C.-M. Su, “Dual-band flat-plate antenna with a shorted parasitic element for laptop applications,” IEEE Trans. Antennas Propag., vol. 53, no. 1, Jan. 2005. [8] K. Chung, T. Yun, and J. Choi, “Wideband CPW-fed monopole antenna with parasitic elements and slots,” IEE Electron. Lett., vol. 40, no. 17, Aug. 2004. [9] H. Shigesawa, M. Tsuji, and A. A. Oliner, “Power leakage from the dominant mode on coplanar waveguides with finite or infinite width,” in 1990 URSI Radio Sci. Meeting Digest, Dallas, TX, May 1990, p. 340. [10] H. Shigesawa, M. Tsuiji, and A. A. Oliner, “Dominant mode power leakage from printed-circuit wave guides,” Radio Sci., vol. 26, pp. 559–564, Mar./Apr. 1991. [11] M. Tsuji, H. Shigesawa, and A. A. Oliner, “New interesting leakage behaviour on coplanar waveguides of finite and infinite widths,” IEEE Trans. Microw. Theory Tech., vol. 39, no. 12, Dec. 1991.

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Rohith K. Raj was born in India. He received the B.Sc. degree in electronics from Mahatma Gandhi University, Kerala, India, and the M.Sc. degree in electronics from P.S.G. College, Coimbatore, India, in 1999 and 2002 respectively. He is currently working towards the Ph.D. degree at Cochin University of Science and Technology, Cochin, India. He is also working as a Project Fellow on the research project “Development of a compact microstrip antenna with reduced radiation hazards suitable for use in mobile communication handsets,” funded by the UGC, Government of India. His current research interest is in compact coplanar printed antenna design and analysis. Mr. Raj was the recipient of an International Union of Radio Science (URSI) Young Scientist award in 2005

K. Vasudevan (SM’84) was born in India. He received the M.Sc. degree in physics from Calicut University, Kerala, India and the Ph.D. degree from Cochin University, Cochin, India, in 1976 and 1982, respectively. From 1980 to 1984, he worked at St. Alberts’ College Ernakulam, Kerala, India. In 1985, he joined the Electronics Department of CUSAT where he is currently the Head of the Department. His research interests includes microstrip antennas, leaky wave antennas and radar cross section studies. Dr. Vasudevan is a Fellow of the Institution of Electronics and Telecommunication Engineers (India).

Manoj Joseph was born in India. He received the B.Sc. degree in physics from Kerala University, Kerala, India, and the M.Sc. degree in electronics science from Cochin University of Science And Technology (CUSAT), Cochin, India, in 2001 and 2003 respectively, where he is currently working towards the Ph.D. degree. His research areas include coplanar antennas, printed monopoles and dipoles. Mr. Joseph was the recipient of an International Union of Radio Science (URSI) Young Scientist award in 2005. He was awarded a Junior Research Fellowship from Kerala State Council for Science, Technology and Environment, Kerala, India.

P. Mohanan (SM’05) was born in India. He received the Ph.D. degree in microwave antennas from Cochin University of Science and Technology (CUSAT), Cochin, India, in 1985. Previously, he worked as an Engineer in the Antenna Research and Development Laboratory, Bharat Electronics, Ghaziabad, India. Currently he is a Professor in the Department of Electronics, CUSAT. He has published more than 100 refereed journal papers and numerous conference articles. He also holds several patents in the areas of antennas and material science. His research areas include microstrip antennas, dielectric resonator antennas, superconducting microwave antennas, reduction of radar cross sections, and polarization agile antennas. Dr. Mohanan received the Career Award from the University Grants Commission in Engineering and Technology, Government of India, in 1994. He is a Reviewer of the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION and IEE Electronics Letters.

C. K. Aanandan was born in India. He received the M.Sc. and Ph.D. degrees from Cochin University of Science and Technology (CUSAT), Cochin, India, in 1981 and 1987, respectively. Currently, he is a Reader in the Department of Electronics, CUSAT. From 1997 to 1998, he worked at the Centro Studi Propagazione E Antenne. Consiglio Nazionale Delle Ricerche, Torino, Italy, under the TRIL program of the International Centre for Theoretical Physics (ICTP). His research interests include microstrip antennas, radar cross section studies and frequency selective surfaces.